Transistor amplifier circuit



Dec. 8, 1953 mc 2,662,124

TRANSISTOR AMPLIFIER CIRCUIT Filed June 1, 1949 5 Sheets-Sheet l 7 F/G.IA ORIENTATION:

I e L/NEAR 4-POLE I l FIG. /8 NOMTION FOR A 7PAN. S/S7'0R.- e I FIG. /C

THE OPEN- CIRCUIT INTERPRE 7717' [0 OF THESE Z 'S GIVES THE FOLLOWINGEQUIVALENT 7':

B. M M/LLA/V 8V PUQQW ATTOPNEV Dec. 8, 1953 B. M MILLAN 2,662,124

TRANSISTOR AMPLIFIER CIRCUIT Filed June 1, 1949 3 Sheets-Sheet 2 M/lE/WUR y B. M M/LLA/V ATTORNEK Dec. 8, 1953 B. MOMILLAN 2,662,124

TRANSISTOR AMPLIFIER CIRCUIT Filed June 1, 1949 5 Sheets-Sheet s BMM/LLA/V. BY

ATTORNEV Patented Dec. 8, 1953 TRANSISTOR AMPLIFIER CIRCUIT BrockwayMcMillan, Summit, N. J., assignor to Bell Telephone Laboratories,Incorporated, New York, N. Y., a corporation of New York ApplicationJune 1, 1949, Serial No. 96,485

3 Claims.

This invention relates in general to electrical translation devices.More particularly, it relates to the amplification and phase inversionof electrical signal currents.

In accordance with the disclosure of application Serial No. 11,165 filedon February 26, 1948, by J. Bardeen and W. 1-1. Brattain, (sinceabandoned in favor of application Serial No. 33,466, filed June 17,1948, which issued on October 3, 1950, as Patent 2,524,035)amplification of electrical signals is obtained by means of a deviceknown as the transistor, which comprises a pair of formed pointelectrodes in rectifier contact with the treated surface of a block ofsemiconducting material such as, for example, germanium, and anadditional electrode in ohmic contact with the body of the block.

In one embodiment of the transistor amplifier as disclosed in Fig. ofPatent 2,524,035 supra, signal currents are impressed between the baseelectrode and ground, i. e., the point of common potential for theexternal circuits to the three electrodes. Itcan be shown that undercertain conditions the output potential appearing on the collectorelectrode of this circuit is reversed in phase with respect to theimpressed input signal potential, thus providing a phaseinvertercircuit. However, because of certain inherent relationships between theinternal and external impedances of the aforesaid circuit, the outputimpedance looking back into the collector circuit is negative under mostconditions, a factor which makes the connection with following circuitsdiflicult. I

Moreover, in many other types of transistor amplifiers disclosed byBardeen-Brattain supra and by others, impedance matching to circuitsconnected to the input and output terminals occasionally presentsproblems because of the large difference between the internal input andoutput impedances of the transistor.

Another characteristic of the transistor amplifier, as embodied in thevarious circuits disclosed by Bardeen-Brattain supra, and by others, isthe inherency of positive feedback through resistance of thesemi-conducting block which is common to the input and output circuits.This factor tends to produce instability under certain operatingconditions.

A principal object of the present invention is to provide improvement inthe operating characteristics of various types of transistor amplifiers.

A more specific object of the present invention is to improve theimpedance-matching char- 2 acteristics of certain types of transistoramplifiers.

Another object of the present invention is to reduce positive feedbackin transistor amplifiers.

Still another object of the present invention is to provide for negativefeedback in transistoramplifier circuits.

These objects and others, which will be apparent hereinafter, arecarried out in a preferred embodiment of the present invention whichcomprises a two-stage phase-inverting amplifier including a pair oftransistors connected in tandem, wherein the emitter of the first stageand the collector of the second stage are connected to ground, 1. e., acommon point of reference potential in the circuit. The collector of thefirst stage is connected for signal transmission to the emitter of thesecond stage. The input signal is impressed on the base electrode of thefirst stage; and the output signal is taken off of the base electrode ofthe second stage.

As will be pointed out in detail hereinafter, such a circuit has certainadvantages, namely, it provides phase inversion of the input signalpotential while presenting a positive impedance looking back into theoutput; it has only a small amount of intrinsic positive feedback; andit has a lower output impedance than a circuit including a singletransistor, with about the same current gain.

The specification describes in detail alternative biasing circuits forthe aforesaid two-stage tandem transistor amplifier.

An additional modification is described for substantially eliminating orneutralizing the residual positive feedback in thetandem amplifierbyintroducing an impedance element which is common to the source andload circuits. If this common impedance element in the tandem circuit issufficiently large, an amplifier is produced having negative feedbackfrom output to input, a feature which is desirable in some applications.

Furthermore, the specification describes, as an independent operatingunit, the second stage of the foregoing two-stage amplifier, including asignal source connected between the emitter electrode and common orground point, and a load impedance connected between the base electrodeand the common or ground point. i

Additional objects and features of the invention will be betterunderstood from a study of the detailed specification and claimshereinafter and the attached drawings of which:

- Figs. 1A-'-1C arediagrams Fig. 2 shows the circuit schematic of oneembodiment of the two-stage tandem phase-inverter transistor amplifierof the present invention including biasing circuits;

Fig. 3 is a simplified schematic including only the signal circuit ofthe tandem amplifier of Fig. 2;

Fig. 4 represents an equivalent circuit of Fig.

Fig. 5 is the circuit schematic of a modified embodiment of Fig. 2showing an alternative arrangement of the biasing circuit;

Fig. 6 shows a modified arrangement .of the signal circuit of Fig. 3 forneutralization of positive feedback;

Fig. '7 shows a subcombination including only the second stage of thetandem amplifier circuit of Fig. 5 connected for independent operatlon;

Fig. 8 is a simplified schematic including only the signal circuit ofthe amplifier of Fig.7; and

Fig. 9 shows an equivalent circuit of "Fig. 8 for theoreticaldiscussion.

Each of the circuits described in the specificationand claimshereinafter includes asits active elements an amplifying device which isknown in the art as a transistor, the construction and operation ofwhich is described in detail in Patent 2,524,035.

'Thebody of the transistor comprises a block of germanium, thecrystalline structure ofwhich is believed to be altered by the presenceof slight quantities of impurities as described in Bardeen- Brattain,supra, to provide different conductivity types, such as, for example,'P-type and N-type. When the major portion of 'the'block comprisesmaterial of one type, for example N-type, the surface of which has beentreated in a manner which is believed to produce a'thin barrier layer ofP-type, the block exhibits remarkable amplifying properties. Formedpoint contacts respectively denoted 'the emitter and the collector, make"rectifying contact with the treated surface of the germanium block. Athird'electrode denoted the base, makes low resistance contact with thebody of the block.

In the specification hereinafter, it has been assumed that'the body ofthe transistor disclosed comprises N-type germanium having a treated orbarrier layer of P-type. However, it is apparent from a study ofBardeen-Brattain, supra, that transistors comprising a block 'havingabody of P-type material witha-barrier layer of N-type material will beequally suitable "for substitution in the circuits describedhereinafter. For the latter case, the polarity of the biases on theemitter 'and collector electrodes will be the "reverse of those for thetransistors described hereinafter.

Asa background for-the discussion hereinafter,

notations and conventions, as applied to transistor circuits, will bediscussed briefly.

Fig. 1A shows a four-terminal device which has two externally accessiblemeshes. It is convenient to describe such devices as four poles, eventhough only two of the three possible external meshes are of interest.

Assuming that currents of the form iIE 'iz are specified arbitrarily inthe two external meshes, where a represents a sinusoidal function 'oftime in terms of s, the natural logarithmic 'base. The term p is thefrequency in radians, .i. e. 21rf (more usually designated w-), and :tis the time in seconds. Then voltages 6.16 e26 appearing across theexternal terminal pairs, are

related to the currents by the following set of equations ez=E21i1+E22izwhere the 2's are complex functions of p.

Equations 1 and 2 are valid under the assumptions that the device'islinear and .the currents i1 and i2 are unrestricted.

Equations 1 and 2 can be symbolized in matrix form as follows:

Placing i2=0, and observing from Equations 1 and' 2the dependence of 61and c2 on i1, it is easy to interpret Z11 as the driving point orselfimpedance of 'mesh I when mesh 2 is open, and Z21 as.the'transimpedance from mesh l to mesh 2 when mesh 2 is open. In asimilar manner, when i1 is placed equal to zero, it may be observed thatZ22 is the self-impedance of mesh 2 when mesh .l

emitter potential, Ie the direct-current emitter current, Vc thedirect-current collector potential, and Is the direct-current collectorcurrent, it has been found that any two of these may be chosen asindependent variables, and the remaining two expressed as functionsthereof.

Adopting Ie, 10 as independent variables, we have the relations:

Ve=Vc(Ie, Ic) (4) Vc=Ve(Ie, I0) (5) Applying small increments AIe, A10to the directcurrent values, one computes the first order incrementsinthe voltages as follows:

From the above, placing 1118:186 Aleige AVe=vee and AVc=Dce theequations take the same form as (1) and (2) above, where:

It is thus apparent that by th choice of current as an independentvariable, the open-circuit impedances are arrived at as parameters fordescribing the linear behavior of the transistor four-pole.

Fig. 10 represents the transistor network of Fig. 1B in the form of anequivalent T network, in which the impedance of the emitter electrode isrepresentedas 2c, of .the collector electrode as ac, .the base electrodeas an, and the net transimpedance as am. The active element of thetransistor .is represented as a voltag generator having polarity asshown, in which the voltage is represented as ant, where i1 is theinstantaneous signal current into the emitter. As indicated in Fig. 10,the impedances of the equivalent transistor circuit can be defined interms of the fourpole impedances developed above:

For the purposes of this discussion, the reactive components of theaforesaid impedances will be neglected, and the resistive componentsthereof will be designated as Te, Tc, Tb, Tm, all of which will beassumed to represent positive values.

For the purpose of the present disclosure and claims, thecurrent'amplification factor a of the transistor may be definedapproximately as the ratio Tm/Tc. The basic transistor terminology thusdefined will now be utilized in a brief theoretical discussion of thecircuit of the present invention.

' Fig. 2 shows cneform of an amplifying device in accordance with thepresent invention comprising two transistors ccnnected in tandemrelation for signal transmission, a source of signal connected to theinput of the tandem pair, a load resistance connected to the output ofthe tandem pair, and associated apparatus.

For proper understanding of the operation of this device, it isconvenient to refer first to Fig. 3 which is a diagrammatic showing ofonly those parts of the circuit which serve to carry appreciablecurrents at signal frequencies, whereas the additional connectionsrequired to supply proper operating voltages to the transistors havebeen omitted. Fig. 3 shows transistor I which is equipped with pointcontact emitter and collector electrodes respectively, 2 and 3,v inrectifying contact with the semiconducting block 4 which comprisesgermanium or similar material} The additional ohmic contact or baseelectrode 5 is attached to the body of the block 4. The transistorelectrodes and semiconducting block ar prepared in a manner described indetail-in the application of Bardeen-Brattain supra. In a similarmanner, the transistor 6 is equipped With emitter, collector and baseelectrodes respectively 1, 8 and 9, incontact with a semiconducting bodyI 0.

In Fig. 3 the emitter 2 of the transistor l and the collector 8 of thetransistor 6 are both connected to the common or ground lead I2 whichrepresents a convenient reference point from which the potentials of theseveral electrodes may b measured. The signal source l3, which maycomprise any conventional signaling circuit, is connected between thebase electrode 5 of the transistor l and the ground point l2, and isrepresented schematically as a generator in series with an impedancewhich in the figure isindicated simply as a resistance Rs. The loadcircuit I4 is connected between the base electrode 9 of the transistorii and the ground point l2, and is represented in the figure simply as aresistance RL. The output of the transistor I, appearing on thecollector electrode 3, isconnected directly to the input of thetransistor 6- through the emitter electrode 1 of the latter.

Fig. 4 shows an equivalent circuit for Fig, 3, with the transistors land 6 indicated as T networks, each having a current generator in thecollector arm. For convenience, the reactive components have beenneglected, and the respective internal transistor impedances areindicated as pure resistances, Te, Tc, Tb and m withappropriatesubscripts. r

The equivalent circuit in Fig. 4 is shown divided by a dotted lineAA.For convenience, co n'- sideration will first be given to that part ofthe circuit to the right of the line AA which is substantially similarto the circuit of Fig. 9, wherein the transistor is shown driven by agenerator l3 of internal impedance Rs. For the purposes of the presentdiscussion, the resistance in the external collector circuit will beassumed equal to zero. 7

, The generator 8a of Fig. 4 represents the active amplification of thetransistor 6, which develops avoltage rm ie with the polarity asindicated, where m is positive. As indicated in Fig. 4;,-the current is,is that flowing through re If the resistance of the external circuit ofcollector "8 is zero or negligible, as stated, a positive curreht'ie,produces a voltage in the generator 8a which tends to bring thepotential of the point X negative relative to the ground point. Ifsufficient current is available from the generator 8a,-the potential ofthe point X may actuallybecome negative despite the current is, which isflowing in such a direction as to make it positive. The condition forthis reversal of polarity is that a, the current amplification of thetransistor, be suiiiciently large. The exact critical value for adepends upon the values of RL, the load resistance, and re the internalemitter resistance of the transistor 6. In any case, this value shouldpreferably exceed unity, but with typical circuit values, need not be asgreat as two. 1 V t The argument just advanced shows that with asufficiently large, the impedance measured between the input terminalsto the second stage of the circuit, that is, the impedance into whichthat part of the circuit to the left of the line AA works, is negative,since a positive applied current is, produces a negative potential atits point of application. This condition will ordinarily obtain withtypical circuit values,

A quite similar argument will show that the circuit to the left of AAalso-presents a negative impedance to the terminals on theline AA; Withthese facts in mind let us now a se sume that in Fig. 4 an instantaneoussignal potential e1 is impressed across the resistance Rs which is highrelative to the internal resistance of the transistor. This produces acurrent ie, in the circuit of the emitter 2 of the transistor whichpasses to ground through the relatively low emitter resistance re, asindicated. Since 7:13 as drawn in Fig. 4, is in the negative direce tionrelative to the orientation conventions of Figs, 1A and 1C, a voltage Tmie with polarity 'opposite to that indicated on the generator 3:;appears across that generator. We have indicated above, however, thatthis generator is in a mesh having a negative impedance. The cur-'- rentie, which flows because of this generator exceed unity. Therefore, undernormal conditions :of: operation, a positive-input voltage: -pro-.--

duces a negative output voltage. That-isthe. outpptpotential-eo. across.the terminals of. the load-,resista-nce "Rn is 'reverseddn phase; withrespecttothe impressed input signal a; which was applied t:-.the baseof.transistor l4. This. .as sumesr that the resistance R1. is largcompared. to the mternal resistances of the. transistor, andthatscertain. mini-mum conditions.- for stability, suclras will. bediscussed hereinafter, havebeenprovided inthe system;

In preferred arrangement, the transistor constants of -.the;.two unitsof the tandem combinatiomare substantially equaL Onthe. basis of an.analysis of. the: symmetrical tandem circuit. ingaccordance with matrixcircuit theory, it. can be; stated thatin. the special case mentionedabove, the current gain not the tandem. unit is essentially equal to thecurrent. gain of a single transistor stage, and the output impedance isapproximately equal to half of that for a single unit, thus: makingv itmore nearly equal to the inputimpedance. V InFig, 2 theloperationforsignal currents is substantially the same as discussed inthe: foregoingparagraph with reference to Fig. 3, and the same. connections areindicated with the exceptionthatinthe emitter circuit 2 of thetransistor I, the connection to the ground point [2 ismad'ethroughacondenser I which has a capacitanceofsucha value as to provide a neg-Iigible. impedance path. for. currents at signal frequenciesuwhileblocking the passage of direct current. Similarly,.the collectorelectrode 8 of thatransistor G'i's connected to ground through thecondenser ls'which performs a like function. Alternative positions I5and 16" for condensers li-andiiFare. indicated in Fig. 2.in dottedlines. Thiiscircuit performs a similar function to the circuitdes'cribed above inblocking. the passage of biasing current through thesignal source. The output. of transistor l is, connected'to the input oftransistor 6 through another condenser llwhich also provides anegligibl'e irnped'ance path for signal currents. and blocks thepassageof'fdirect current.

For the purpose of supplying proper operating or bias potentials,direct-current. sources are providedto maintain the electrodesrespectively positiveand negative with reference to the base electrode,as provided in I the disclosurev of Bardeen-.Brattain supra. Theseinclude. the..posis tive direct-current biasing circuit for. the.emitter. electrode Z'ofthe transistor Iiwliich includes the.directcurrentsource lT' connected in series with, the resistanceelement22't0'th'ebase 616C: trod'e 5. The negative biasing circuit for thecollector 3'of the transistor l includes the direct; current source l8which is connected in series with the resistance. element; 23 to-thebase; elec-.- trode 5f In a similar manner, the emitter. T and thecollector 8v of the transistor 6' are respectively biased positively andnegatively by the directcurrent source I9 in series with the, resistanceelement 2'4. and the direct-current source. in serieswith the resistanceelement. 251.

For the,purposesof discussion, the respective potentialsof'thebiasingsources I1, 18, I9 and 20. willbe, designated E0 .Ec Eg andv Ecand the resistancevalues of the respective resistance elements 22, 23,24 and 25' will be designated.Rs,, R1,,R2 and R1.,. In allinstances. thesourcesof potential, which may be conventionalbatteries, are connectedbetweenpoints acrosswhichan of current at-signal frequencies.

appreciable potential existsmt signal frequencies.- during: operation oftherdeviceas ampl fier. Accordingly,v in Fig." 2,. resistancesfl, 23, 24and 2.5-.are,provided series with the. respective po. tential sourcesl1, l8 l.9. and 20 to provide pas. sage for the direct current necessaryto support proper bias potentials while at the same time offeringrelatively high resistance paths to currents at signal frequencies; Moregenerally, any of these resistors couldbe replaced by a twoterminarnetwork having such properties that: it presents a; closed path 1 to.the passage oidirect current, preferably at relatively; low impedance;whileaofieringr a high-impedance. to the: passage To provideproper-biasing voltages, account must be: taken of ;the. voltage. drop=in'these blocking. resistors .or networks; caused by: the flow of biascurrent. Typical values will be discussed subsequently.

amplifying devices employing transistors; one may encountenthe questionof stability, ins asmuch as an improper choice of circuitvalues mayvcause the system. to go into self-oscillation. Thev complete study ofthe stabilityoi anampliflerwhich, operates over a wide range offrequencies iscomplex, and cannot be-entered ill-i302 here. The-methodsoutlined in the book, Network. Analysis, and Feedback Amplifier De-.sign, by H. W. Bode, D- Van llostrand Co. 194 5, are available forthispurpose. It is characteristic otmost devices employing transistors,however, thatthey-will be. stable, provided the source and load.circuits are of sufficiently high impedance. A sufficient condition forthe stability of the simplified circuit of, Fig, 3, forexample, isthatRsand Robe such that.

(Rs-PRU) (RH-R22) R12R21 (8) stability. of: the: device;

Returning to the physicaldevice as; exhibited in'iFig: 2;..certainzfurther conditions are imposed becauseaot theadditionalcurrentpaths intro.- duced'pbycthezbias circuits. In the'first place; ashas alreadybeenmentioned, these current paths are; in shunt; with 1certain of' the desired signal paths indicated in Fir; 3; Therlossofsignal cure rent: in: these-shuntipaths;carrbe made small. by making.these pathsiotsnmciently high impedancei-Typical-Wellies:torenderthesezlosses small might .he; 2.55 follows:

Es -#1000 ohms Er -50,000 ohms. 121 1000 ohms- RL, 50,000- ohms Inaddition to these loss considerations; .the stability characteristics ofthe system again come into question. Rs, is effectively in parallel withRs atsignal frequencies and 'RL, in parallel with a... The relation (8)abovev for'R's andR'tmust then be modified by replacing these quantitiesrespectively by s s RLRL Rs-i-Rs RL+ R1.1 In addition, the R11, R22, R12and R21 of that relation now depend not only on the transistorcharacteristics but also upon the values chosen for R1 and R2. Thisdependence is easily computed, but yields a formula too complex to haveany illustrative value. With the values suggested above for R1 and R2,in typical narrow band systems, the influence of these resistors can beneglected. In wide band systems the analytic methods described in thebook of H. W. Bode cited above, may be of assistance in consideringquestions relating to stability.

The circuit of Fig. 5, in which the transistors and their respectiveelectrodes, the signal source and the load are connected in the samemanner as corresponding units in Fig. 2, and bear the same designations,shows a second possible form of the device differing from that in Fig. 2in the arrangement of the bias circuits. In Fig. 5,

the emitter 2 of the transistor I, and the collector s 8 of thetransistor 6 are biased by the respective direct-current sources I1 and2t, which are connected directly to ground. The collector 3 of thetransistor I is biased by a circuit which includes the biasing resistor23, and the directcurrent source l8, connected directly to ground; andthe emitter I of the transistor 6 is biased by a similar circuit whichincludes the biasing resistor 24 and the direct-current source l9.

This arrangement may have certain advantages in that all bias sourcesnow appear at ground potential. For the purposes of discussion, assumethat the potentials of the direct-current sources l1, l8, I9 and 20' arerespectively designated as E6 E01, Ee and En and that the values ofresistance elements 23' and 24' are designated as R1 and R2.

In Fig. the bias circuits are somewhat interlocked, but the necessarybiasing source voltages can be calculated readily from the followingdirect-current relations:

Let

with additional subscripts 1 and 2 to designate the correspondingtransistor. Then with the sign conventions of the battery voltages Es E0etc., as shown on the diagram Typical values of the bias potentials andcurrents are Vc=+1 volt, I=+.5 ma., Vc=+3O volts, Ic=3 ma. Thedependence of the stability of the system upon R1 and R2 is now morecritical. A choice of RiZlOQOOO ohms and RzilOODOO ohms will, however,render their influence negligible, in typical narrow band systems.

In accordance with one of the features of the invention, as describedhereinbefore, the amplified signal current appearing in the load M isopposite in phase to the signal voltage appearing across the sourcegenerator is. This feature provides a relatively simple means ofneutralizing or eliminating the feedback which leads to the instabilityof the circuit discussed above. The schematic diagram of Fig. 6 showshow this can be done. In this figure the emitter 2 of the transistor iand the collector 8 of the transistor 6 are connected to a common pointwhich is separated from the ground point [2 by a resistor 26' having avalue RF. It will be noted that the resistance RF has been introduced asan impedance common to meshes which carry both source and load currents.Because of the aforementioned phase-reversing property of the device,the signal current in RF introduces a voltage in the source mesheffectively in series with the source generator, which is opposite inphase to that introduced by the source generator. Therefore RF producesnegative feedback. A choice of RF=R12, where R12 is the circuittransimpedance in a reverse direction referred to in the relation (8)hereinbefore, is such as to make this negative feedback exactly cancelthe intrinsic positive feedback of the device. This cancellation rendersthe system stable independently of the choice of Rs and RL- As mentionedabove, a typical value might be RF= ohms. An increase of RF beyond thevalue R12 introduces further negative feedback which may be desirable insome applications. More generally, and in wide band systems, theresistor RF would be replaced by a two-terminal network whosecharacteristic wouldbe one of the parameters used in designing thesystem to achieve the desired stability and transmissioncharacteristics.

The schematic of Fig. 6 may be completed to includes the bias sources ineither of the ways indicated in Figs. 2 and 5. A simple modification ofthe direct-current bias relations stated for Fig. 5 is required toaccount for the voltage drop in RF- Consider now, the characteristics ofthe second stage of the tandem combination of Fig. 3 operated as anamplifier, independently of the first stage, a signal source connectedin series with the emitter electrode, and the load connected in serieswith the base electrode.

Fig. 7 shows an amplifying device comprising the aforesaid combinationwhich includes a transistor 6, a source I 3 of signal voltage, a load H3and associated circuits. The transistor 6 is the same as the transistor6 described hereinbefore with reference to Figs. 2'through 6, andcomprises an emitter electrode 1, a collector electrode 8 and a baseelectrode 9 all in contact with a semiconducting block I 8. Fig. 8 showsthe essential components of Fig. 7, without the auxiliary circuitsrequired to supply operating voltages to the transistor. In Fig. 8, thecollector electrode 8 of the transistor 6 is connected to the common orground point !2 through a network 33 shown simply as a resistor R. Thesource 13, shown as a generator with self-impedance Rs, is connectedbetween ground l2 and the emitter electrode 1. The load l l, shown as aresistor RL, connects be tween the base electrode 9 and ground 12. Theseare the paths for signal currents.

Fig. 9 shows an equivalent circuit at signal frequencies for theamplifying device of Figs. 7 and 8.

As shown with respect to the second stage of Fig. 4, when a, the currentgain, is sufliciently large, the impedance measured between the inputterminals to the equivalent circuit of Fig. 9-that is, the impedanceinto which the source [3 works-is negative, since a positive appliedcurrent 2'9. produces a negative potential at its point of application,a condition which ordinarily obtains with typical circuit values.

The existence of a negative impedance at the .Design, by H. W. Bode.

[11 input'of the :device :places a restrictiononi the value of: Rs inorder that the system be stable. It is evident that if Rs Were-papositive impedance 'of such a value asrexactlytoicancelithetnegativeinput impedance across which it is connected, the

mesh in which is flows would have zero net simpedance and wouldtherefore oscillate. As mentioned hereinbefore, for the full analysis ofthe stability of the system in wide bandapplications,

methods may be'employedsuch as those described in the book, NetworkAnalysis and Feedback In narrow band systems,.it will suffice in,general to choose. Rs,'JRr. and R to satisfy the following condition inorder to insure stability:

(RS-i-R-t-Te-I-Tc-Tm) (RL+R+Tc+Tb) g In this relation, r n, Tb and Tmare the constants of the transistoras indicated in Fig. 9. Typicalvalues of these constants might be r='200 ohms Tb=100 ohms Tc=5,000 ohmsT1n=10,-000 ohms With these values, -and.Rs 10,000 ohms, ,RL O,

R=0, the device wouldbe stable. Increasing the value of Rintroducesnegative feedback, which may. in-itself'be desirable in someapplications,

.and at the same time relaxes the restrictionsron Rs and R1. as imposedby the inequality set forth in Equation 13.

:Returningnow tov Fig. 7, the current .path from the source [3 to theemitterelectrodei is completed through a condenser 29 which offers a lowimpedance path for currents at signal frequencies but blocks the passageof direct current. A potential source-30, which may be -a battery havinga -potentialEe, is connected to the emitter I through a resistor 3i tosupplydirect-current bias. The resistor 3|, having a resistance valuewhich will be designated Rs provides-passage for the .biasingcurrentwhile presenting a high-impedance to currents atsignal. frequencies..Aipotential source-32, which may be in the. form of a .batteryhaving apotential E connected .in the lead of the collector electrode 8 providesbias voltage fonthat electrode.

. In .Fig. 7, the only additional conditions imposed by the presence ofthe bias circuits are:

(i) The resistor Rs represents a .shunt path in which signal current islost. To reduce this loss, the impedance of ,Rs rto. currents: at signalfre- -quencies should be kept high; a value RSFZKLOOO ohmswould beadequate in typical circuits.

(ii) Looking from-the input terminals of the transistor, Rs, is,efiectively inparallel with -:Rs

at signal'frequencies and the relation (13) ,above must therefore bemodified by replacing Re by V R R Rs-t- Rs More generally '(inparticular, in wide band applications), all. of the resistors Rs, R51,R1,. and

"R should preferably be replaced by two terminal :networks whoseimpedance functions would be parameters under control of the designer.The design methods outlined in the book of HJW.

Bode referred to above would then be applied to produce a system whosestability, feedback-and transmission properties were suitable to theintended application. A detailed discussion of these design problemsinithe present-specification would be superfluous.

12 I'o'dnsuraproper .biascpotentialsiiin 'the: circuit of Fig. 7, thefollowing direct-current'rrelations are recommended.

Let

Ve=desired emitter voltageabove base .V=desired' collectori'voltage"below .base

.le resulting emitter current I=resulting collector current -'I'henwiththe sign iconventionshof the .battery voltages Ee, Es, shown on the.diagram Ee=(Ie-Ic) .RL+'IeRS +Ve 4) E0: (IeIc) RL+IR+V0 (15) Typicalvalues of the bias "potentials and currentsare:

'Ve==+ 1 volt I e=+ .5 ma.

Vc=+30 volts IFS .ma.

"It is apparentthat the concepts bfthe present "invention can be carriedout in other'embodiments than those specifically shown and de-'scr'ibe'd herein for the-purposes of illustration.

What is claimed is: I 1.' In" combination, a first semiconductor body,

a-second semiconductor body,:eachof said bodies having emitter,collector and base electrodes 00- operatively associated-therewith,means for connecting a source of signals between the -base electrode ofthe first-body and 'a'poin't of'reference potential, means forconnecting the emitter electrode of'the first body through an externalsignal transduc'ing connection to said point ofreferencepotentia1,-means for coupling the collector electrode ofthe-'first-body tothe emitter electrode of the secon'dbodyin "signaltransfer relation, means'for connecting the'col- .for connecting auseful load circuit between-the baserelectrode of thesecondof'saidtransistors and said point of reference potential, the collectorelectrode of thefirst said transistor being coupled in signal transfer:relation to the emitter electrode of the second said transistor, two

sources of biasing current each having a negligible signal impedance'tosaid point of reference potential, one of said sources-connected betweensaid point of reference potential and the emitter electrode of saidfirst stage and theother of said sources connected between said point ofreference potential and the collector electrode of said secondstage,-two additional sources of biasing current for respectivelysupplying biasing current to the collector electrode of said firsttransistor and the emitter electrode of said second transistor, and'apair of impedance elements, each of said last-named biasing-sourcesconnected between its respective electrode and midpoint of referencepotential in series with one of said impedance elements,

13 3. A circuit in accordance with claim 2 in which the parallelcombination of said impedance elements has a value substantially inexcess of the input impedance of the transistor comprised in the secondstage of said amplifier.

BROCKWAY McMILLAlQ'.

References Cited in the file of this patent UNITED STATES PATENTS NumberName Date 1,846,043 Terman Feb. 23, 1932 2,067,048 Gill et al. Jan. 5,1937 2,101,699 Baesecke et a1 Dec. 7, 1937 14 2,431,333 Labin Nov.25,1947 6.3 3 Rack July 19, 1949 2,486,776 Barney Nov. 1, 19 9 2,517,960Barney Aug. 8, 1950 2,524,035 Bardeen et a1 Oct. 3, 1950 2,533,001Eberhard Dec. 5, 1950 2,541,322 Barney Feb. 13, 1951 2,569,347 ShockleySept. 25, 1951 2,585,078 Barney Feb. 12, 1952 OTHER REFERENCES RadioEngineering, text by Terman, 3d ed., pp. 308-311, published 1947 byMcGraw-Hill Book Co., New York. (Copy in Div.

